Feedback-Based Linearization of Voltage Controlled Oscillator

ABSTRACT

Embodiments of the present invention enable a feedback-based VCO linearization technique. Embodiments include a frequency locked loop formed by feeding back a VCO&#39;s output into the VCO&#39;s input in negative phase by means of a frequency-to-voltage (F/V) converter. Embodiments enable constant VCO gain over a wide input tuning range and across PVT variations. Further, embodiments can be nested within a PLL, for example, with negligible area and power consumption overhead.

BACKGROUND

1. Field of the Invention

The present invention relates generally to linearization of voltagecontroller oscillators (VCOs).

2. Background Art

Conventionally, VCOs use MOSFET variable capacitors (varactors) and thusare inherently non-linear.

Various calibration methods exist for linearizing VCOs. For example, onetechnique attempts to operate the VCO in a linear region of its gainresponse function. However, this technique only works for small signalmodulation and does not work when modulating wide bandwidth signals,such as WCDMA and EDGE signals, for example. Another technique uses aset of varactors biased in a staggered fashion so as to generate in theaggregate a linear transient response of the VCO. However, thistechnique does not address VCO gain variations due to process, voltage,and temperature (PVT) variations. In addition, because each varactorrequires a clean reference voltage for operation, implementation of thistechnique is both difficult and expensive.

Thus, conventional VCO gain linearization techniques are not suitablefor generating complex wide bandwidth waveforms, cannot handle PVTvariations, and are relatively difficult and expensive to implement.

Accordingly, there is a need for improved methods and systems forlinearizing the gain of a VCO.

BRIEF SUMMARY

Embodiments of the present invention relate generally to linearizationof voltage controller oscillators.

Embodiments of the present invention, as will be further describedbelow, enable a feedback-based VCO linearization technique. Embodimentsinclude a frequency locked loop formed by feeding back a VCO's outputinto the VCO's input in negative phase by means of afrequency-to-voltage (F/V) converter. Embodiments enable constant VCOgain over a wide input tuning range and across PVT variations. Further,embodiments can be nested within a PLL, for example, with negligiblearea and power consumption overhead.

Further embodiments, features, and advantages of the present invention,as well as the structure and operation of the various embodiments of thepresent invention, are described in detail below with reference to theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

The accompanying drawings, which are incorporated herein and form a partof the specification, illustrate the present invention and, togetherwith the description, further serve to explain the principles of theinvention and to enable a person skilled in the pertinent art to makeand use the invention.

FIG. 1 illustrates an example frequency locked loop (FLL) forlinearizing a voltage controlled oscillator (VCO) according to anembodiment of the present invention.

FIG. 2 illustrates a phase locked loop (PLL) having a nested FLLaccording to an embodiment of the present invention.

FIG. 3 illustrates an example switched capacitor circuit.

FIG. 4 illustrates an example FLL according to an embodiment of thepresent invention.

FIG. 5 illustrates another example FLL according to an embodiment of thepresent invention.

FIG. 6 illustrates an example implementation of a portion of a FLLaccording to an embodiment of the present invention.

FIG. 7 illustrates an example implementation of a non-overlapping clockgenerator circuit according to an embodiment of the present invention.

FIG. 8 illustrates an example implementation of a frequency to voltage(F/V) converter according to an embodiment of the present invention.

FIG. 9 illustrates an example implementation of a filter circuitaccording to an embodiment of the present invention.

FIG. 10 illustrates an example implementation of an offset currentsource circuit according to an embodiment of the present invention.

FIG. 11 illustrates an example implementation of a variable resistoraccording to an embodiment of the present invention.

FIG. 12 illustrates an example implementation of a variable capacitoraccording to an embodiment of the present invention.

FIG. 13 illustrates an example implementation of an integrator circuitaccording to an embodiment of the present invention.

The present invention will be described with reference to theaccompanying drawings. Generally, the drawing in which an element firstappears is typically indicated by the leftmost digit(s) in thecorresponding reference number.

DETAILED DESCRIPTION OF EMBODIMENTS

Linear gain voltage controlled oscillators (VCOs) are desirable in manyapplications, including phase locked loop (PLL)-based phase modulatorsand frequency synthesizers, for example. In phase modulators, it isdesired that the VCO gain is as linear as possible to not disturb theoutput of the phase modulator. For example, a non-linear gain VCO maycause unwanted frequency components to appear at the output of the phasemodulator, possibly violating the transmission mask and/or the adjacentchannel leakage ratio (ACLR) specifications. In frequency synthesizers,because the transient response and the loop bandwidth of the frequencysynthesizer are strongly dependent on the gain of the VCO (K_(VCO)),non-linearities in the VCO gain cause the loop bandwidth of thefrequency synthesizer to vary with frequency, resulting in stabilityproblems.

Thus, generally, constant VCO gain is desired over a wide range of inputtuning voltage. In addition, constant VCO gain over process, voltage,and temperature (PVT) variations is desired as it simplifiessignificantly the design of PLL-based phase modulators and frequencysynthesizers. For example, constant VCO gain over PVT variationseliminates the need for VCO calibration, simplifying the complexity ofany on-chip DSP engines and reducing the time required for startupadjustments.

Conventionally, VCOs use MOSFET variable capacitors (varactors) and thusare inherently non-linear. Various calibration methods exist forlinearizing VCOs. For example, one technique attempts to operate the VCOin a linear region of its gain response function. However, thistechnique only works for small signal modulation and does not work whenmodulating wide bandwidth signals, such as WCDMA and EDGE, for example.Another technique uses a set of varactors biased in a staggered fashionso as to generate in the aggregate a linear transient response of theVCO. However, this technique does not address VCO gain variations due toPVT variations. In addition, because each varactor requires a cleanreference voltage for operation, implementation of this technique isboth difficult and expensive.

Thus, conventional VCO gain linearization techniques are not suitablefor generating complex wide bandwidth waveforms, cannot handle PVTvariations, and are relatively difficult and expensive to implement.

Accordingly, there is a need for improved methods and systems forlinearizing the gain of a VCO.

Embodiments of the present invention, as will be further describedbelow, enable a feedback-based VCO linearization technique. Embodimentsinclude a frequency locked loop formed by feeding back a VCO's outputinto the VCO's input in negative phase by means of afrequency-to-voltage (F/V) converter. Embodiments enable constant VCOgain over a wide input tuning range and across PVT variations. Further,embodiments can be nested within a PLL, for example, with negligiblearea and power consumption overhead.

FIG. 1 illustrates an example frequency locked loop (FLL) 100 forlinearizing a voltage controlled oscillator (VCO) according to anembodiment of the present invention.

As shown in FIG. 1, FLL 100 includes a VCO 102, a F/V converter 104, anda subtractor 106. An input tuning voltage V_(TUNE) 108 is coupled to thepositive terminal of subtractor 106, and an output frequency signalf_(OUT) 110 is generated at the output of VCO 102. A feedback loopincluding F/V converter 104 couples output frequency signal f_(OUT) 110to the negative terminal of subtractor 106.

Assuming that the gains of VCO 102 and F/V converter 104 are K_(VCO) andK_(F2V) respectively, the gain F_(VCO−FB) of FLL 100 can be written as:

$\begin{matrix}{K_{{VCO}\text{-}{FB}} = {\frac{K_{VCO}}{1 + {K_{VCO} \cdot K_{F\; 2V}}}.}} & (1)\end{matrix}$

With K_(VCO.)K_(F2V)>>1, equation (1) can be approximated as:

$\begin{matrix}{K_{{VCO}\text{-}{FB}} = {\frac{1}{K_{F\; 2V}}.}} & (2)\end{matrix}$

Thus, effectively, the gain of FLL 100 is independent of the gainK_(VCO) of VCO 102 and only depends on the gain K_(F2V) of F/V converter104. As a result, FLL 100 can be made linear and PVT independent bydesigning F/V converter 104 to have constant and PVT independent gain.

According to embodiments, FLL 100 can be nested within a PLL, in aPLL-based phase modulator or frequency synthesizer, for example. FIG. 2illustrates an example embodiment having FLL 100 nested in a PLL. It isnoted that in such embodiment it is necessary that FLL 100 has a highenough bandwidth (i.e., speed) compared to the PLL, in order not todisturb the PLL output. However, when this bandwidth requirement issatisfied, the PLL response function continues to have the samecharacteristics as with an open-loop VCO, but with K_(VCO) replaced with1/K_(F2V) in the response function.

As noted above, FLL 100 can be made linear and PVT independent bydesigning F/V converter 104 to have constant and PVT independent gain.In practice, a F/V converter is equivalent to a FM (frequencymodulation) detector or demodulator. Conventional FM detectors (e.g.,slope detector, Foster-Seely discriminator, ratio detector, gated-beamdetector, etc.) are all based on the dependence of an inductor'sreactance on frequency. As such, besides being bulky and unsuitable forintegration into a PLL, conventional F/V converters exhibit a lineargain dependence on frequency. Thus, conventional F/V converters cannothave constant gain across a wide input frequency range.

Embodiments of the present invention, as further described below, employa switched capacitor circuit to produce a linear and PVT independent F/Vconverter. Since switched capacitor circuits are inexpensive and smallin size, embodiments can be easily integrated into a PLL, for example.

FIG. 3 illustrates an example switched capacitor circuit 300 accordingto an embodiment of the present invention. As shown in FIG. 3, switchedcapacitor circuit 300 includes a capacitor 308 and two switches 312 and314. Switches 312 and 314 are controlled respectively by control signalsCLK 316 and CLK 318, which are non-overlapping clock signals offrequency f_(CLK). A reference voltage V_(REF) 306 is coupled at inputnode A 302 of switched capacitor circuit 300. The output voltage,V_(OUT), of switched capacitor circuit 300 is measured across a resistor310, coupled between output node B 304 and ground.

Because CLK 316 and CLK 318 are non-overlapping clock signals, capacitor308 is alternately coupled to input node A 302 and output node B 304 ata switching frequency equal to f_(CLK). Thus, at every switching cycle,capacitor 308 transfers a charge from V_(REF) 306 to resistor 310 at theswitching frequency f_(CLK).

It can be shown that an effective resistance, R_(SW), between input node302 and output node 304 is equal to 1/(C_(SW).f_(CLK)), where C_(SW) isthe capacitance of capacitor 308. Therefore, when the resistance R ofresistor 310 is significantly lower than R_(SW), the output voltageV_(OUT) can be written as:

$\begin{matrix}\begin{matrix}{V_{OUT} = {i_{OUT} \cdot R}} \\{\approx {\frac{V_{ref}}{R_{S\; W}} \cdot R}} \\{= {V_{ref} \cdot C_{S\; W} \cdot f_{CLK} \cdot R}} \\{= {\left( {V_{ref} \cdot R \cdot C_{S\; W}} \right) \cdot f_{CLK}}} \\{= {k_{F\; 2V} \cdot f_{{CLK}\;}}}\end{matrix} & (3)\end{matrix}$

Accordingly, the output voltage, V_(OUT), of switched capacitor circuit300 is a function of switching frequency f_(CLK). Furthermore, it isnoted that the gain k_(F2V) of switched capacitor circuit 300 isindependent of frequency and is a function of V_(REF) (a constant) andthe time constant R.C_(SW). Thus, if the time constant R.C_(SW) can becalibrated for PVT variations (which can be achieved using simpleon-chip RC calibration circuitry), switched capacitor circuit 300provides a linear, PVT independent frequency to voltage converter.

FIG. 4 illustrates an example FLL 400 according to an embodiment of thepresent invention. As shown in FIG. 4, FLL 400 includes in itsfeed-forward section an operational amplifier-based integrator 402followed by VCO 102. In its feedback loop, FLL 400 includes a frequencydivider 404, F/V converter 300, and a filter 406.

Integrator 402 is an embodiment of subtractor 106, described above inFIG. 1. Thus, integrator 402 subtracts the voltage output of filter 406from input tuning voltage V_(TUNE) 108. The resulting error voltagesignal is input into VCO 102.

In an embodiment, as shown in FIG. 2, VCO 102 is a differential outputoscillator. Thus, VCO 102 generates frequency signals Out+ 408 and Out−410, which are fed back through frequency divider 404 to F/V converter300. In an embodiment, frequency divider 404 divides-down frequencysignals Out+408 and Out− 410 by a pre-determined integer n, and furtherincludes clock generating circuitry for generating non-overlapping clocksignals 316 and 318, described above with reference to FIG. 3. Inanother embodiment, a separate clock generating circuitry, coupledbetween frequency divider 404 and F/V converter 300, is used to generateclock signals 316 and 318.

F/V converter 300, as described above with reference to FIG. 3, receivesnon-overlapping clock signals 316 and 318 and generates an outputvoltage representative of the frequency of clock signals 316 and 318.

Filter 406 filters out the output voltage of F/V converter 300. In anembodiment, filter 406 is a current-mode second-order low-pass filter.It is noted that the low-pass nature of the feedback loop of FLL 100helps clean the phase noise of VCO 102 at close-in offset frequencies.This is in addition to the fact that the feedback loop of FLL 100 helpsreduce the phase noise of VCO 102 within the bandwidth of FLL 100.

It is noted that from equation (2) above that the output frequency ofFLL 100 can be written as:

$\begin{matrix}{f_{OUT} = {{\left( \frac{1}{K_{F\; 2V}} \right) \cdot V_{TUNE}} = {\left( \frac{1}{V_{ref} \cdot R \cdot C_{S\; W}} \right) \cdot V_{TUNE}}}} & (4)\end{matrix}$

Thus, the output frequency of FLL 100 is a linear function of the inputtuning voltage, insensitive to non-linearities inherent in VCO 102, PVTvariations, and frequency of operation. Moreover, with a simple on-chipRC calibration circuit and a constant reference voltage, the value ofthe gain (or the sensitivity) of FLL 100 can be set very accurately.

FIG. 5 illustrates another example FLL 500 according to an embodiment ofthe present invention. Example FLL 500 is similar to example FLL 400described above with reference to FIG. 4. In addition, as shown in FIG.5, FLL 500 includes an offset current source 504, coupled at the outputof filter 406. Offset current source 504 compensates for the inherentoffset in VCO 102. In an embodiment, offset current source 504 absorbsthe output current of filter 406 when VCO 102 is free running (forV_(TUNE) equal to zero).

FLL 500 also includes a load resistor R_(L) 502 coupled at the output offilter 406. Resistor 502 corresponds to resistor 310, described above inFIG. 3. Thus, resistor 502 and capacitor 308 provide the time constantof switched capacitor circuit 300. In an embodiment, as furtherdescribed below, resistor 502 and capacitor 308 are variable so thattheir RC time constant can be calibrated for PVT variations. Inaddition, by being variable, resistor 502 and capacitor 308 enable thegain of FLL 500 to be tunable as desired.

Furthermore, FIG. 5 shows an example implementation of filter 406, whichincludes a current mirror circuit (comprised of elements 506 and 510 inFIG. 5) and a low-pass filter 508. In an embodiment, low-pass filter 508is characterized by two real poles. It is noted that the current mirrorcircuit allows filter 406 to be insensitive to non-linearities and PVTvariations. In particular, because elements 506 and 510 have inversetransconductance gains, any non-linearities and PVT variations in filter406 are eliminated.

Example implementations of various portions of FLL embodiments of thepresent invention are provided below. Embodiments of the presentinvention are not limited to the example implementations providedherein, but extend to any other implementations, variations, orimprovements that would be apparent to a person skilled in the art basedon the teachings herein.

FIG. 6 illustrates an example implementation 600 of a portion of a FLLaccording to an embodiment of the present invention. In particular,example implementation 600 shows a portion of the feed-forward sectionof the FLL and a portion of the feedback loop of the FLL. In thefeed-forward section, example implementation 600 includes VCO 102,coupled to a buffer 602, and a divide-by-2 divider 604. An open-draindriver circuit 606 is coupled at the output of the FLL to provide adifferential output of the FLL. In the feedback loop, exampleimplementation 600 includes a buffer 608, followed by two divide-by-2dividers 610 and 612 and a non-overlapping clock generator 614.

Dividers 604, 610 and 612 perform in the aggregate a division by 8 ofthe output of VCO 102 to generate inputs 616 and 618 of clock generator614.

Clock generator 614 generates non-overlapping clock signals CLK 316 andCLK 318 based on inputs 616 and 618.

FIG. 7 illustrates an example implementation 700 of a non-overlappingclock generator circuit according to an embodiment of the presentinvention. Example clock generator circuit 700 may be an implementationof non-overlapping clock generator 614, for example. As shown in FIG. 7,example clock generator circuit 700 receives an input clock signal ClkIn702 and generates as output non-overlapping clock signals CLK 316 andCLK 318. In an embodiment, clock generator circuit 700 includes twoparallel branches (will be referred to hereinafter as upper and lowerbranches), as shown in FIG. 7. The upper branch includes an inverter704, followed by a NOR gate 706 and inverters 708 and 710. The lowerbranch includes a buffer 712, followed by a NOR gate 714 and inverters716 and 718. NOR gates 706 and 714 are cross-coupled, as shown in FIG.7, via multiplexers 720 and 722. In an embodiment, multiplexers 720 and722 are 4:1 multiplexers.

FIG. 8 illustrates an example implementation 800 of a frequency tovoltage (F/V) converter according to an embodiment of the presentinvention. Example F/V converter 800 may be an implementation of F/Vconverter 104, for example. Example F/V converter 800 is similar toexample F/V converter 300 described above with reference to FIG. 3.However, F/V converter 800 uses a differential switched capacitorcircuit implementation, instead of the single-ended implementation ofF/V converter 300. Thus, as shown in FIG. 8, F/V converter 800 includestwo pairs of switches 802 and 804 (implemented in an embodiment usingNMOS transistors), with switches 802 a and 802 b controlled by clocksignal CLK 316 and switches 804 a and 804 b controlled by clock signalCLK 318. In addition, F/V converter 800 generates output signals 806 and808. Operation of F/V converter 800 is similar to the operation of F/Vconverter 300, described above with reference to FIG. 3.

FIG. 9 illustrates an example implementation 900 of a filter circuitaccording to an embodiment of the present invention. Example filtercircuit 900 may be an implementation of filter 406, for example. Asshown in FIG. 9, example filter 900 adopts a differential-to-singleended topology, receiving input signals 902 and 904 and generatingoutput signal 906. In an embodiment, input signals 902 and 904correspond respectively to output signals 806 and 808 of example F/Vconverter 800, described above. It is noted that by having adifferential input, the need to stabilize the quiescent point of theinput of example filter 900 against PVT variations is alleviated.Further, the size of the capacitors within example filter 900 can bereduced.

FIG. 10 illustrates an example implementation 1000 of an offset currentsource circuit according to an embodiment of the present invention.Example current source circuit 1000 may be an implementation of offsetcurrent source 504, for example.

FIG. 11 illustrates an example implementation 1100 of a variableresistor according to an embodiment of the present invention. Examplevariable resistor 1100 may be an implementation of variable loadresistor 502, for example.

FIG. 12 illustrates an example implementation 1200 of a variablecapacitor according to an embodiment of the present invention. Examplevariable capacitor 1200 may be an implementation of switching capacitor308, for example.

FIG. 13 illustrates an example implementation 1300 of an integratorcircuit according to an embodiment of the present invention. Exampleintegrator circuit 1300 may be an implementation of integrator 402, forexample.

It is to be appreciated that the Detailed Description section, and notthe Summary and Abstract sections, is intended to be used to interpretthe claims. The Summary and Abstract sections may set forth one or morebut not all exemplary embodiments of the present invention ascontemplated by the inventor(s), and thus, are not intended to limit thepresent invention and the appended claims in any way.

Embodiments of the present invention have been described above with theaid of functional building blocks illustrating the implementation ofspecified functions and relationships thereof. The boundaries of thesefunctional building blocks have been arbitrarily defined herein for theconvenience of the description. Alternate boundaries can be defined solong as the specified functions and relationships thereof areappropriately performed.

The foregoing description of the specific embodiments will so fullyreveal the general nature of the invention that others can, by applyingknowledge within the skill of the art, readily modify and/or adapt forvarious applications such specific embodiments, without undueexperimentation, without departing from the general concept of thepresent invention. Therefore, such adaptations and modifications areintended to be within the meaning and range of equivalents of thedisclosed embodiments, based on the teaching and guidance presentedherein. It is to be understood that the phraseology or terminologyherein is for the purpose of description and not of limitation, suchthat the terminology or phraseology of the present specification is tobe interpreted by the skilled artisan in light of the teachings andguidance.

The breadth and scope of embodiments of the present invention should notbe limited by any of the above-described exemplary embodiments, butshould be defined only in accordance with the following claims and theirequivalents.

What is claimed is:
 1. A frequency locked loop (FLL) having afeed-forward section and a feedback section, comprising: in thefeed-forward section, a voltage controlled oscillator (VCO) thatreceives a control voltage signal and generates an output frequencysignal; in the feedback section, a frequency-to-voltage (F/V) converterthat receives the output frequency signal and generates a feedbackvoltage signal; and in the feed-forward section, a subtractor thatreceives an input tuning voltage and the feedback voltage signal andgenerates the control voltage signal; wherein a gain of the FLL isindependent of a gain of the VCO.
 2. The FLL of claim 1, wherein theoutput frequency signal of the VCO is a linear function of the inputtuning voltage.
 3. The FLL of claim 1, wherein the gain of the FLL islinear and independent of process, voltage, and temperature (PVT)variations.
 4. The FLL of claim 1, wherein the gain of the FLL isindependent of frequency.
 5. The FLL of claim 1, wherein the gain of theFLL only depends on a gain of the F/V converter.
 6. The FLL of claim 5,wherein the F/V converter comprises a switched capacitor circuit havinga variable switching capacitor and a variable load resistor, and whereinthe gain of the F/V converter is a function of a RC time constant of theswitched capacitor circuit.
 7. The FLL of claim 6, wherein the RC timeconstant of the switched capacitor circuit is substantially constant. 8.The FLL of claim 6, further comprising calibration circuitry coupled tothe switched capacitor circuit.
 9. The FLL of claim 8, wherein thecalibration circuitry calibrates the variable switching capacitor andthe variable load resistor to maintain the RC time constant of theswitched capacitor circuit at a fixed value independent of process,voltage, and temperature (PVT) variations.
 10. The FLL of claim 8,wherein the calibration circuitry adjusts the variable switchingcapacitor and the variable load resistor to vary the gain of the F/Vconverter and the FLL.
 11. The FLL of claim 6, wherein the switchedcapacitor circuit further comprises a plurality of switches that receivenon-overlapping control clock signals and that alternately couple thevariable switching capacitor to a reference voltage supply and thevariable load resistor, thereby transferring charge from the referencevoltage supply to the variable load resistor at a switching frequency ofthe clock signals.
 12. The FLL of claim 11, wherein an output voltage ofthe switched capacitor circuit across the variable load resistor is afunction of the switching frequency of the non-overlapping control clocksignals.
 13. The FLL of claim 1, wherein the subtractor comprises anoperational amplifier-based integrator circuit.
 14. The FLL of claim 1,further comprising in the feedback section: one or more frequencydividers coupled between the VCO and the F/V converter; and a filter,coupled to an output of the F/V converter, that filters the feedbackvoltage signal.
 15. The FLL of claim 14, wherein the filter includes acurrent-mode second-order low-pass filter.
 16. The FLL of claim 14,wherein the filter includes a current mirror circuit that enables thefilter to be insensitive to non-linearities and process, temperature,and voltage (PVT) variations.
 17. The FLL of claim 14, furthercomprising in the feedback section: an offset current source, coupled toan output of the filter, that absorbs an output current signal of thefilter when the input tuning voltage is equal to zero.
 18. The FLL ofclaim 1, further comprising in the feedback section: a non-overlappingclock generator circuit, coupled between the VCO and the F/V converter,that receives the output frequency signal and that generatesnon-overlapping control clock signals.
 19. The FLL of claim 18, whereinthe F/V converter receives the non-overlapping control clock signals andgenerates the feedback voltage signal as representative of a frequencyof the non-overlapping control clock signals.
 20. The FLL of claim 1,wherein the VCO is a differential output oscillator, wherein thefrequency output signal comprises differential frequency output signals.21. The FLL of claim 1, wherein the FLL is nested within a phase lockedloop (PLL).